The present invention relates to a speed control apparatus for an electric motor of the type utilizing alternative current in the armature, and more specifically to a variable speed control apparatus for a synchronous motor of the type having a revolving magnetic field.
The synchronous motor of the type having a rotary magnetic field is provided with an armature defining a stator and field poles defining a rotor. Multi-phase alternative current flowing in the stator windings of the motor generate revolving magnetic fields which cause the rotor to synchronously rotate by magnetic pulling forces acting on it. It is helpful for understanding of the invention to explain here the relationships of torques to be generated between a direct current (D.C) motor and a synchronous motor, prior to the description of the conventional speed control apparatus for the synchronous motor.
FIG. 1A illustrates a sectional view on the rotation axis of a D.C. motor, wherein FM represents field poles, AM is an armature, AW is armature windings, and DCV is a direct current voltage source. FIG. 1B illustrates Fleming's left hand law concerning armature current (Ia), field or magnetic flux (.phi.) and torque (T) to be generated on the D.C. motor shown in FIG. 1A. As shown in FIGS. 1A and 1B, a rectifier RC switches armature current Ia so that it always flows in a direction perpendicular to that of magnetic flux .phi.. In the case, this torque T to be generated is as follows: EQU T=K.multidot..phi..multidot.Ia (1)
where K is a constant. The equation (1) shows that torque T is proportional to armature current Ia under the condition that the magnitude of magnetic flux .phi. is kept constant.
FIG. 2 illustrates a sectional view on the rotation axis of a synchronous motor provided with rotary field magnetic poles PM, wherein "SW" designates stator windings and "Is" designates the current vector flowing in the stator windings "SW". In order to apply the equation (1) to a synchronous motor provided with rotary field magnetic poles as shown in FIG. 2, it is necessary to correspond the magnetic flux .phi. to the magnetic flux vector .phi.s of field poles PM, and further to correspond armature current Ia to the current vector Is of the stator windings SW. Accordingly, the torque T to be generated in the synchronous motor is given in the following equation (2): EQU T=K.multidot..phi.s.multidot.Is.multidot.cos .gamma. (2)
FIG. 3 illustrates an electric circuit equivalent to the synchronous motor shown in FIG. 2. In the figure, Ra designates an electric resistance of the stator (armature) windings, and Xs designates a reactance equivalent to the reaction and the leakage magnetic flux on the stator windings. Further, V is a voltage source, and .gamma. is the phase difference between the armature current Is and the electromotive force Eo induced on the stator windings by the armature current. Therefore, when the phase difference .gamma. is zero, that is, the electromotive force Eo is in phase with the armature current Is, the equation (2) becomes the following: EQU T=K.multidot..phi.s.multidot.Is (3)
The equation (3) shows the possibility that the synchronous motor can be driven in the same fashion as in torque generation with a D.C. motor. In other words, the equation (3) indicates that it is necessary for efficiently driving the synchronous motor to control the armature current Is flowing in the stator windings so as to make the revolving field magnetic flux .phi.s always at a right angle with the armature current Is.
Furthermore, the armature current Is is defined as a function of first order lag to the rotary angular velocity .omega. of the synchronous motor, which includes a time constant Ta formed with electric resistance Ra and inductance La of the stator windings. Accordingly, it results that the phase lag of the armature current Is is produced as the rotation speed of the synchronous motor becomes high. This is a conventional way to adopt a current control loop in the drive unit of the synchronous motor for making the phase lag smaller.
FIG. 4 illustrates control block diagrams corresponding to the drive unit of the motor. In the figure, Gv designates a gain of a current amplifier in the unit, and Gi designates a current feedback gain in the unit.
In FIG. 4, the transfer function G(s) of the unit is defined as follows: ##EQU1## where S=j.omega.. Furthermore, open loop gain Go(s) is defined as follows: EQU Go(S)=Gi.multidot.Gv/Ra(1+STa) (5)
When Go(S)&gt;&gt;1, however, the transfer function G(S) becomes as follows: EQU G(S)=1/Gi (6)
The equation (6) indicates that the transfer function G(ds) is independent of the time constant Ta and rotary angular velocity .omega. under the condition of Go(S)&gt;&gt;1. This is the reason why the current control loop is provided on the drive unit of the synchronous motor.
FIG. 5 illustrates a conventional type of control circuit for drive control of a synchronous motor. In the figure, reference numeral 2 designates a two phase resolver for producing a phase modulated signal with respect to a rotary angle .theta. of a rotor 7 mounted on a synchronous motor 6, and reference numerals 2a, 2b are primary windings of resolver 2, each being COS windings and SIN windings to which sinusoidal wave signals cos (.omega.ot) and sin (.omega.ot) are applied from a reference signal generator 4, respectively. Reference numeral 2c designates a secondary winding mounted on a rotor of resolver 2, the rotor being fixed on rotor 7 of motor 6. A signal Sin (.omega.ot+.theta.) is produced on secondary winding 2c as electromotive force generated between primary winding 2a 2b and secondary windings 2c.
Reference numeral 8 designates a multiplier for multiplying the output signal sin (.omega.ot+.theta.) and an output signal V of a velocity instruction unit 10. Reference numeral 12 designates a wave shaper for shaping an output signal sin (.omega.ot) into rectangular pulse trains.
Reference numeral 14 designates a synchronous rectifier circuit (or phase descriminating circuit) for synchronously rectifying an output signal V.multidot.sin (.omega.ot+.theta.) of multiplier 8 with the rectangular wave signal from wave shaper 12. Reference numeral 16 designates a current control loop for producing a control signal I* sin .theta. applied to a known power drive unit 18 formed with semiconductor devices. The current control loop 16 receives the signal V.multidot.sin .theta. and a current feed back signal I.sub.fb which is produced at a detecting unit 17 which detects a current actually flowing in the armature of motor 6. Reference numeral 20 designates an alternative power source.
FIGS. 6A, 6B are detailed circuit block diagrams of synchronous rectifier 14 shown in FIG. 5. FIGS. 6A and 6B illustrate digital and analogue types of synchronous rectifiers, respectively.
As shown in FIG. 6A, a digital type of synchronous rectifier 14A is provided with a wave shaper 22 for forming a rectangular wave signal 23 from the signal sin (.omega.+.theta.), a differentiator 24 for producing a pulse signal Pr by differentiating in time an output signal 21 of wave shaper 12 at a time corresponding to the rising up of each output rectangular signal 21 as shown in FIG. 6C, another differentiator 26 for producing a pulse signal Pr' by differentiating in time output signal 23 of wave shaper 22 at a time corresponding to rising up of each signal 23 as shown in FIG. 6C, and a flip flop circuit 28 which is set by the pulse Pr and reset by the pulse Pr'. The synchronous rectifier 14A is further provided with an AND gate 32 which allows clock pulses Cp from a pulse generator 30 to pass when flip flop 28 is set, a counter 34 for counting the number of the clock pulses supplied from AND gate 32, and a converter 36 for producing a signal V.multidot.sin .theta. from the values of counter 34 and speed instruction signal V, the value of counter 34 corresponding to the phase difference .theta. between the reference signal sin .omega.ot and a phase modulated signal sin (.omega.ot+.theta.) by resolver 2. Converter 36 includes a reference memory table 36A for referring to the values .theta. and converting them to sin .theta. for processing in a digital to analogue converter 36B. A multiplier 35 for multiplying the speed instruction signal V and the signal sin .theta. is also provided.
FIG. 6B illustrates an analogue type of synchronous rectifier 14. In the figure, reference numeral 38 designates a wave shaper whose output signal is logically inverted in an inverter 45 and applied to a switch 44 in analogue switch unit 40. Another output signal of wave shaper 38 is applied to a switch 42 in analogue switch unit 40. Switch 42 allows the signal V.multidot.sin (.omega.ot+.theta.) to pass to a low pass filter 48 which forms a signal V.multidot.sin .theta.. Switch 44 allows inverted signal 47 of V.multidot.sin (.omega.ot+.theta.), which is formed by an analogue inverter unit 41 having an operational amplifier 46, an input resistance 46A and a feedback resistance 46B, to pass to low pass filter 48.
FIG. 7A illustrates a circuit diagram of current control loop 16 shown in FIG. 5. In the figure, reference numeral 16A designates an operational amplifier having gain Gi. Reference numerals 16B, 16C and 16D are electric resistances. Reference numeral 16E is a capacitor.
Hereinafter, the functions and operations of circuit block diagrams shown in FIG. 5 will be explained.
In the figure, when rotor 7 rotatably mounted on synchronous motor 6 rotates at its angular velocity .omega. and its rotary angle becomes at an angle .theta. as shown in the figure, secondary winding 2C of resolver 2 produces a phase modulated signal sin (.omega.ot+.theta.). Multiplier 8 multiplies speed instruction signal V and the phase modulated signal sin (.omega.ot+.theta.), thereby producing the signal V.multidot.sin (.omega.ot+.theta.). Synchronous rectifier 14 synchronously rectifies the signal V.multidot.sin (.omega.ot+.theta.) by a rectangular wave signal given from wave shaper 12 and produces the signal V.multidot.sin .theta. corresponding to the angular position of rotor 7. This signal V.multidot.sin .theta. and current feedback signal I.sub.fb are supplied to current control loop unit 16 which produces the control signal I* sin .theta. applied to the semiconductor type drive unit 18. In the unit 18, the signal I* sin .theta. is modified into multi-phase alternative current signals, each phase signal being supplied to power inverting devices corresponding to each armature winding mounted on the stator of motor 6. Thus, each armature current makes a revolving magnetic field in motor 6. As mentioned above, since each armature current is supplied in accordance with the rotor' s rotation angle .theta. equal to the direction of magnetic field in the motor, the phase difference between the directions of the magnetic flux .theta. and each armature current can be held at a right angle .pi./2. However, as the rotation speed becomes high and the angular velocity .omega. becomes large, the equation (5) does not hold.
FIG. 7B shows a transfer function G(j.omega.) which has frequency characteristics depending on the values of ".omega.". In the figure, the output signal I.sub.a sin (.theta.-.phi.) includes a phase lag .phi. because of large values of the rotary angular velocity .omega.. In this case, .phi. is defined as the phase angle of the transfer function G(j.omega.), as follows: EQU .phi.=arg.multidot.G(j.omega.)
FIG. 7C is a general control block diagram for suggesting a basic technical concept for eliminating phase lag .phi. indicated in FIG. 7B. In the figure 7C, a block 40 suggests a possibility of phase lag elimination by the way that the signal V.multidot.sin .theta. is operated with .phi. formed by the motor's rotation speed v and a constant k, thereby advancing the phase .theta. by the angle .phi..
Furthermore, as shown in FIG. 4 and FIG. 5, block element Gv corresponding to a power amplifier in drive unit 18 has, in general, some delay circuit elements, thereby the total phase lag depending on the rotary angular velocity .omega. becomes that much larger. Therefore, in the conventional speed control apparatus for the synchronous motor shown in FIG. 5, a desired controlability cannot be obtained when the motor rotates at high speed because phase differences between the revolving magnetic field and armature current (Ia) are generated.